1. Field of the Invention
The present invention relates to a circuit for the efficient operation of switched reluctance machines, particularly those machines which are required to be operated in both motoring and generating modes.
2. Description of Related Art
In general, a reluctance machine is an electrical machine in which torque is produced by the tendency of its movable part to move into a position where the reluctance of a magnetic circuit is minimized, i.e. where the inductance of the exciting winding is maximized. In some forms, circuitry is provided for detecting the angular position of the rotor and energizing the phase windings as a function of the rotor position. This type of reluctance machine is generally known as a switched reluctance machine and it may be operated as a motor or a generator. The characteristics of such switched reluctance machines are well known and are described in, for example, “The Characteristics, Design and Application of Switched Reluctance Motors and Drives” by Stephenson and Blake, PCIM'93, Nürnberg, Jun. 21-24, 1993, incorporated herein by reference. That paper describes in some detail the features of the switched reluctance machine which together produce the characteristic cyclically varying inductance of the phase windings. It is well-known in the art that such machines can be operated in either the motoring or generating mode simply by altering the timing of the winding excitation, without altering the direction of current flowing in the phase winding(s).
FIG. 1 shows the principal components of a typical switched reluctance drive system, connected to a load 19. The input DC power supply 11 can be either a battery or rectified and filtered AC mains and can be fixed or variable in magnitude. In some known drives, the power supply 11 includes a resonant circuit which produces a DC voltage which rapidly varies between zero and a predetermined value to allow zero voltage switching of power switches. The DC voltage provided by the power supply 11 is switched across the phase windings 16 of the motor 12 by a power converter 13 under the control of the electronic control unit 14. The switching must be correctly synchronized to the angle of rotation of the rotor for proper operation of the drive. A rotor position detector 15 or a position detection algorithm supplies signals indicating the angular position of the rotor. Some form of current transducer 17 is normally provided to give phase current feedback. The input demand 18 can be a torque or speed demand for motoring or a current or voltage demand for generating.
The energization of the phase windings in a switched reluctance machine depends on detection of the angular position of the rotor. This may be explained by reference to FIGS. 2 and 3, which illustrate the switching of a reluctance machine operating as a motor. FIG. 2 generally shows a rotor pole 20 approaching a stator pole 21 according to arrow 22. As illustrated in FIG. 2, a portion 23 of a complete phase winding 16 is wound around the stator pole 21. When the portion 23 of the phase winding 16 around stator pole 21 is energized, a force will be exerted on the rotor, tending to pull rotor pole 20 into alignment with stator pole 21. FIG. 3 generally shows typical switching circuitry in the power converter 13 that controls the energization of the phase winding 16, including the portion 23 around stator pole 21 of stator 24. The voltage busses 36, 37 are generally known as the DC link and the capacitor 35 across them is known as the DC link capacitor, whose function is to handle the alternating currents on the DC link. When switches 31 and 32 are closed, the phase winding is coupled to the source of DC power and is energized. When the phase winding of a switched reluctance machine is energized in the manner described above, the magnetic field set up by the flux in the magnetic circuit gives rise to the circumferential forces which, as described, act to pull the rotor poles into line with the stator poles.
In general, the phase winding is energized to effect the rotation of the rotor as follows. At a first angular position of the rotor (called the “turn-on angle”, θon), the controller 14 provides switching signals to turn on both switching devices 31 and 32. When the switching devices 31 and 32 are on, the phase winding is coupled to the DC link, causing an increasing magnetic flux to be established in the machine. The magnetic flux produces a magnetic field in the air gap which acts on the rotor poles to produce the motoring torque. The magnetic flux in the machine is supported by the magneto-motive force (mmf) which is provided by a current flowing from the DC supply 11 through the switches 31 and 32 and the phase winding 23. Current feedback is generally employed and the magnitude of the phase current is controlled by chopping the current by rapidly switching one or both of switching devices 31 and/or 32 on and off. FIG. 4(a) shows a typical current waveform in the chopping mode of operation, where the current is chopped between two fixed levels. In motoring operation, the turn-on angle θon is often chosen to be the rotor position where the center-line of an inter-polar space on the rotor is aligned with the center-line of a stator pole, but may be some other angle. FIG. 4(a) also shows the form of the idealized inductance profile of the phase winding.
In many systems, the phase winding remains connected to the DC link (or connected intermittently if chopping is employed) until the rotor rotates such that it reaches what is referred to as the “freewheeling angle”, θfw. When the rotor reaches an angular position corresponding to the freewheeling angle (e.g., the position shown in FIG. 2) one of the switches, for example 31, is turned off. Consequently, the current flowing through the phase winding will continue to flow, but will now flow through only one of the switches (in this example 32) and through only one of the diodes 33/34 (in this example 34). During the freewheeling period, the voltage drop across the phase winding is small, and the flux remains substantially constant. The circuit remains in this freewheeling condition until the rotor rotates to an angular position known as the “turn-off angle”, θoff, (e.g. when the center-line of the rotor pole is aligned with that of the stator pole). When the rotor reaches the turn-off angle, both switches 31 and 32 are turned off and the current in phase winding 23 begins to flow through diodes 33 and 34. The diodes 33 and 34 then apply the DC voltage from the DC link in the opposite sense, causing the magnetic flux in the machine (and therefore the phase current) to decrease.
It is known in the art to use other switching angles and other current control regimes. Similarly, many other configurations of lamination geometry, winding topology and switching circuitry are known in the art, some of which are discussed in the Stephenson & Blake paper cited above.
As the speed of the machine rises, there is less time for the current to rise to the chopping level, and the drive is normally run in a “single-pulse” mode of operation. In this mode, the turn-on, freewheel and turn-off angles are chosen as a function of, for example, speed and load torque. Some systems do not use an angular period of freewheeling, i.e. switches 31 and 32 are switched on and off simultaneously. FIG. 4(b) shows a typical such single-pulse current waveform where the freewheel angle is zero. It is well known that the values of turn-on, freewheel and turn-off angles can be predetermined and stored in some suitable format for retrieval by the control system as required, or can be calculated or deduced in real time.
FIG. 5(a) shows the current and flux waveforms for a typical single-pulse operating point in motoring. It will be noted that the current and flux begin to rise as soon as the voltage is applied to the phase winding at θon. The voltage is applied for a conduction angle θc. At the switch-off angle θoff, the gradient of the flux reverses as the voltage across the coil is reversed and, in the absence of eddy currents in the magnetic circuit, the current and flux reach zero together. This Figure is drawn for the ideal case of zero resistance, so the rise and fall times are equal. FIG. 5(b) shows the waveforms for the corresponding generating point, in which the waveforms are a mirror image in time of those of FIG. 5(a).
It will be noted that the currents in the switches and in the diodes are generally different. For example, in the motoring mode of FIG. 5(a) the switch current is significantly greater than the diode current—indeed the root mean square (rms) value is probably three times that of the diode current. By contrast, in the generating mode of FIG. 5(b) the diode current has the larger value. It follows that if the drive is required to provide broadly the same output in motoring and generating the ratings of the switches and diodes must be broadly the same.
The losses associated with the currents in the switches and diodes may be significant, either with respect to the cooling of the devices themselves, to keep them within their thermal operating limits, or with respect to the overall efficiency of the drive. In an industrial drive operating from, say, a 400V 3-phase supply and developing tens or hundreds of kilowatts, the device losses will probably be below 3% of the total input power. In a low-voltage drive however, the devices losses can be 10% or more of the input.
One solution to this problem is to use devices with larger silicon area, and this will improve the situation to a limited extent, albeit at the expense of the size and cost of the devices. However, this is beneficial mainly to the switches, since the voltage drop characteristic of the diodes is either a constant value (e.g. 1.0V) or a constant plus a current dependent term (e.g. 0.5V+0.05i). This is not too much of a problem in the motoring mode, when the diode current is relatively small. However, in the generating mode the larger current often produces losses which are a serious drawback to efficient operation.
It would be possible to replace the diodes with thyristors, as shown in FIG. 6, but there is no advantage in this, for two reasons. Firstly, the timing of the switching of the thyristor would be absolutely critical, since the thyristor would need to be fired at the instant the corresponding main switch was opened. Because of finite switching times associated with the devices (generally of the order of 10-20 μsec, depending on its size), it would be extremely difficult to ensure correct firing at all times. Secondly, even if the first objection were overcome, the voltage drop across the thyristor would be greater than across the diode it replaced, leading to larger, not smaller, device losses.
Other types of switch would be precluded, for the same or additional reasons. For example, an Integrated Gate Bipolar Transistor (IGBT) could be considered on the grounds that it would have a smaller device loss than a thyristor, but replacing the diode by an IGBT would not work, since the reverse voltage across the IGBT would destroy it.
It has been known to use standard configurations of IGBTs and diodes, which are available in industry-standard modules, for a switched reluctance drive. For example, the paper “Starter-alternator for hybrid electric vehicle: Comparison of induction and variable reluctance machines and drives” by Miller et al, Proceedings of the 1998 IEEE Industry Applications Society Annual Meeting, Oct. 12-15, 1998, St Louis, Mo., Vol. 1, pp 513-523, incorporated herein by reference, describes the arrangement of IGBTs and diodes reproduced in FIG. 7. The use of this circuit is described in the paper at p 517, column 2. In one cycle of phase conduction, the IGBTs SA & SD are used as switches and the return diodes are DB and DC. IGBTs SB and SC are biased to the off condition. In the following cycle of phase conduction, the IGBTs SC & SB are used as switches and the return diodes are DA and DD. IGBTs SA and SD are biased to the off condition. This is done so that current will flow in opposite directions in the phase winding on successive cycles, thus reducing mutual coupling between adjacent phases, as described at page 518, column 1, lower half. While this achieves the high torque output sought by the authors, it is a very costly solution, requiring eight devices per phase.
European Patent Application No. 0886370, incorporated herein by reference, also discloses a switching circuit in which a half bridge module is connected between the voltage rails. A diode is connected across each switch to conduct towards the positive rail. The phase current reverses direction on alternate cycles or on groups of cycles.